The present invention relates to a switching power supply circuit including a voltage resonant converter.
As so-called soft-switching power supply of a resonant type, a current resonant type and a voltage resonant type are widely known. In a present situation, a current resonant converter having two switching devices coupled by a half-bridge coupling system is in wide use because such a current resonant converter is easily put to practical use.
However, the characteristics of a high withstand voltage switching device, for example, are now being improved, and therefore the problem of withstand voltage in putting a voltage resonant converter to practical use is being cleared up. In addition, a voltage resonant converter formed by a single-ended system with one switching device is known to be advantageous as compared with a current resonant forward converter having one switching device in terms of input feedback noise, the noise component of a direct-current output voltage line, and the like.
FIG. 12 shows an example of configuration of a switching power supply circuit including a voltage resonant converter of the single-ended system.
In the switching power supply circuit shown in FIG. 12, a rectifying and smoothing circuit formed by a bridge rectifier circuit Di and a smoothing capacitor Ci rectifies and smoothes an alternating input voltage VAC, and thereby generates a rectified and smoothed voltage Ei as a voltage across the smoothing capacitor Ci.
Incidentally, a noise filter formed by a set of common mode choke coils CMC and two across capacitors CL and removing common-mode noise is provided in the line of a commercial alternating-current power supply AC.
The rectified and smoothed voltage Ei is input as a direct-current input voltage to the voltage resonant converter. As described above, the voltage resonant converter employs the single-ended system with one switching device Q1. The voltage resonant converter in this case is an externally excited converter. The MOS-FET switching device Q1 is switching-driven by an oscillation and drive circuit 2.
A MOS-FET body diode DD is connected in parallel with the switching device Q1. A primary-side parallel resonant capacitor Cr is connected in parallel with the source and drain of the switching device Q1.
The primary-side parallel resonant capacitor Cr forms a primary side parallel resonant circuit (voltage resonant circuit) in conjunction with the leakage inductance L1 of a primary winding N1 of an isolated converter transformer PIT. This primary side parallel resonant circuit provides a voltage resonant operation as the switching operation of the switching device Q1.
The oscillation and drive circuit 2 applies a gate voltage as a drive signal to the gate of the switching device Q1 to switching-drive the switching device Q1. Thus the switching device Q1 performs switching operation at a switching frequency corresponding to the cycle of the drive signal.
The isolated converter transformer PIT transmits the switching output of the switching device Q1 to a secondary side.
The isolated converter transformer PIT has for example an EE type core formed by combining E-type cores of ferrite material with each other. A winding part is divided into a primary side winding part and a secondary side winding part. The primary winding N1 and a secondary winding N2 are wound around the central magnetic leg of the EE type core.
In addition, a gap of about 1.0 mm is formed in the central magnetic leg of the EE type core of the isolated converter transformer PIT. Thereby a coupling coefficient k=about 0.80 to 0.85 is obtained between the primary side and the secondary side. The coupling coefficient k at this level may be considered to represent loose coupling, and therefore a state of saturation is not easily obtained. The value of the coupling coefficient k is a factor in setting the leakage inductance (L1).
One end of the primary winding N1 of the isolated converter transformer PIT is inserted between the switching device Q1 and the positive electrode terminal of the smoothing capacitor Ci. Thereby, the switching output of the switching device Q1 is transmitted to the primary winding N1. An alternating voltage induced by the primary winding N1 occurs in the secondary winding N2 of the isolated converter transformer PIT.
In this case, a secondary side parallel resonant capacitor C2 is connected in parallel with the secondary winding N2. Thus, the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonant capacitor C2 form a secondary side parallel resonant circuit (voltage resonant circuit).
In addition, a half-wave rectifier circuit is formed by connecting a rectifier diode Do1 and a smoothing capacitor Co to the secondary side parallel resonant circuit as shown in the figure. This half-wave rectifier circuit generates a secondary side direct-current output voltage Eo having a level corresponding to once an alternating voltage V2 obtained in the secondary winding N2 (secondary side parallel resonant circuit) as a voltage across the smoothing capacitor Co. The secondary side direct-current output voltage Eo is supplied to a load, and is also input to a control circuit 1 as a detection voltage for constant-voltage control.
The control circuit 1 inputs a detection output obtained by detecting the level of the secondary side direct-current output voltage Eo input as the detection voltage to an oscillation and drive circuit 2.
The oscillation and drive circuit 2 controls the switching operation of the switching device Q1 according to the level of the secondary side direct-current output voltage Eo which level is indicated by the detection output input to the oscillation and drive circuit 2 so as to make the secondary side direct-current output voltage Eo constant at a predetermined level. That is, the oscillation and drive circuit 2 generates and outputs a drive signal for controlling the switching operation. Thereby control is performed to stabilize the secondary side direct-current output voltage Eo.
FIGS. 13A and 13B and FIG. 14 show results of experiments on the power supply circuit having the configuration shown in FIG. 12. In conducting the experiments, principal parts of the power supply circuit of FIG. 12 are set as follows as conditions of VAC=100 V corresponding to an AC 100 V system.
For the isolated converter transformer PIT, an EER-35 core is selected, and the gap of the central magnetic leg is set to a gap length of 1 mm. As for the respective numbers of turns of the primary winding. N1 and the secondary winding N2, N1=43 T and N2=43 T. As for the coupling coefficient k of the isolated converter transformer PIT, k=0.81 is set.
The primary-side parallel resonant capacitor Cr=6800 pF and the secondary side parallel resonant capacitor C2=0.01 μF are selected. Accordingly, the resonant frequency fo1=175 kHz of the primary side parallel resonant circuit and the resonant frequency fo2=164 kHz of the secondary side parallel resonant circuit are set.
The rated level of the secondary side direct-current output voltage Eo is 135 V. Load power handled by the power supply circuit is in a range of maximum load power Pomax=200 W to minimum load power Pomin=0 W.
FIGS. 13A and 13B are waveform charts showing the operations of principal parts in the power supply circuit shown in FIG. 12 on the basis of the switching cycle of the switching device Q1. FIG. 13A shows a switching voltage V1, a switching current IQ1, a primary winding current I1, a secondary winding voltage V2, a secondary winding current I2, and a secondary side rectified current ID1 at the maximum load power Pomax=200 W. FIG. 13B shows the switching voltage V1, the switching current IQ1, the primary winding current I1, the secondary winding voltage V2, the secondary winding current I2, and the secondary side rectified current ID1 at the minimum load power Pomin=0 W.
The switching voltage V1 is a voltage obtained across the switching device Q1. The voltage V1 is at a zero level in a period TON in which the switching device Q1 is on, and forms a resonant pulse having a sinusoidal waveform in a period TOFF in which the switching device Q1 is off. The resonant pulse waveform of the voltage V1 indicates that the operation of the primary side switching converter is voltage resonant type operation.
The switching current IQ1 flows through the switching device Q1 (and the body diode DD). The switching current IQ1 is at a zero level in the period TOFF. In the period TON, the switching current IQ1 flows through the body diode DD in a forward direction and is thus of negative polarity at a turn-on time, and is thereafter inverted to flow through the drain and the source of the switching device Q1 and increase until turn-off timing. Therefore, the switching current IQ1 has a peak level in the turn-off timing.
The primary winding current I1 flowing through the primary winding N1 is obtained by combining a current component flowing as the switching current IQ1 in the period TON with a current flowing through the primary-side parallel resonant capacitor Cr in the period TOFF. The primary winding current I1 has a waveform as shown in the figures.
As for the operation of a secondary side rectifier circuit, at the maximum load power Pomax=200 W, the rectified current ID1 flowing through the rectifier diode Do1 has a peak level at the time of turning on the rectifier diode Do1, and thereafter decreases to a zero level in a waveform as shown in FIG. 13A. The rectified current ID1 is at the zero level during the off period of the rectifier diode Do1. Incidentally, at the minimum load power Pomin=0 W, the rectified current ID1 is at the zero level even during the on period.
The secondary winding voltage V2 in this case is obtained in the parallel circuit of the secondary winding N2 and the secondary side parallel resonant capacitor C2. The secondary winding voltage V2 is clamped at the secondary side direct-current output voltage Eo during the on period during which the secondary side rectifier diode Do1 conducts. During the off period of the secondary side rectifier diode Do1, the secondary winding voltage V2 forms a sinusoidal waveform in a direction of negative polarity. The secondary winding current I2 flowing through the secondary winding N2 is obtained by combining the rectified current ID1 with a current flowing through the secondary side parallel resonant circuit (N2(L2)//C2). The secondary winding current I2 flows in waveforms as shown in FIGS. 13A and 13B, for example.
FIG. 14 shows switching frequency fs, the on period TON and the off period TOFF of the switching device Q1, and AC→DC power conversion efficiency (ηAC→DC) with respect to load variation in the power supply circuit shown in FIG. 12.
The AC→DC power conversion efficiency (ηAC→DC) is 90% or more in a range of load power Po=100 W to 200 W. It is known that the single-ended system in which the voltage resonant converter has one switching device Q1, in particular, provides favorable power conversion efficiency.
The switching frequency fs, the on period TON, and the off period TOFF shown in FIG. 14 represent switching operation as characteristics of constant-voltage control against load variation in the power supply circuit of FIG. 12. In this case, the switching frequency fs is controlled to be raised as the load becomes lighter. As for the on period TON and the off period TOFF, the off period TOFF is substantially constant irrespective of load variation, whereas the on period TON is shortened as the load becomes lighter. That is, the power supply circuit shown in FIG. 12 variably controls the switching frequency in such a manner as to reduce the on period TON as the load becomes lighter, for example, while keeping the off period TOFF constant.
By thus variably controlling the switching frequency, inductive impedance obtained by providing the primary side parallel resonant circuit and the secondary side parallel resonant circuit is varied. By varying the inductive impedance, an amount of power transmitted from the primary side to the secondary side and an amount of power transmitted from the secondary side parallel resonant circuit to the load are changed, so that the level of the secondary side direct-current output voltage Eo is changed. The secondary side direct-current output voltage Eo is thereby stabilized.
FIG. 15 schematically shows the constant-voltage control characteristics of the power supply circuit shown in FIG. 12 by relation between the switching frequency fs (kHz) and the secondary side direct-current output voltage Eo.
Letting fo1 be the resonant frequency of the primary side parallel resonant circuit and fo2 be the resonant frequency of the secondary side parallel resonant circuit, in the circuit of FIG. 12, the secondary side parallel resonance frequency fo2 is lower than the primary side parallel resonance frequency fo1, as described above.
As for constant-voltage control characteristics with respect to the switching frequency fs under a condition of a constant alternating input voltage VAC, as shown in FIG. 15, characteristic curves A and B respectively represent constant-voltage control characteristics at the maximum load power Pomax and the minimum load power Pomin under the resonant impedance corresponding to the resonant frequency fo1 of the primary side parallel resonant circuit, and characteristic curves C and D respectively represent constant-voltage control characteristics at the maximum load power Pomax and the minimum load power Pomin under the resonant impedance corresponding to the resonant frequency fo2 of the secondary side parallel resonant circuit.
Further, when the primary side parallel resonant circuit and the secondary side parallel resonant circuit are provided as in the circuit of FIG. 12, there is an intermediate resonant frequency fo between the resonant frequencies fo1 and fo2. A characteristic curve E represents a resonant impedance characteristic based on relation between the intermediate resonant frequency fo and the switching frequency fs at the maximum load power Pomax. A characteristic curve F represents a resonant impedance characteristic based on the relation between the intermediate resonant frequency fo and the switching frequency fs at the minimum load power Pomin.
With the voltage resonant converter provided with the secondary side parallel resonant circuit, the level of the secondary side direct-current output voltage Eo is determined by the resonant impedance characteristics of the intermediate resonant frequency fo in relation to the switching frequency fs. The voltage resonant converter shown in FIG. 12 employs a so-called lower side control system in which the switching frequency fs is variably controlled in a frequency region lower than the intermediate resonant frequency fo.
A variable range (necessary control range) of the switching frequency fs which range is necessary to achieve constant voltage with the rated level (135 V in the case of the circuit of FIG. 12) of the secondary side direct-current output voltage Eo as a target value by switching frequency control corresponding to lower side control under the characteristics represented as the characteristic curves E and F corresponding to the intermediate resonant frequency fo in FIG. 15 is a section indicated by Δfs. In other words, by varying the switching frequency to a required value according to load variation in the frequency range corresponding to the section indicated by Δfs, the secondary side direct-current output voltage Eo is controlled to be at the rated level tg.
For details, reference should be made to Japanese Patent Laid-open No. 2000-152617.
With diversification of various electronic apparatuses, there is a demand for a so-called wide range capability that enables the power supply circuit to operate dealing with the commercial alternating-current power supply input of either of an AC 100 V system and an AC 200 V system.
As described above, the power supply circuit having the configuration shown in FIG. 12 operates so as to stabilize the secondary side direct-current output voltage Eo by switching frequency control, and the variable range (necessary control range) of the switching frequency fs which range is necessary to stabilize the secondary side direct-current output voltage Eo is indicated by Δfs as described with reference to FIG. 15.
The power supply circuit shown in FIG. 12 deals with load variation in a relatively wide range of 200 W to 0 W. The actual necessary control range of the power supply circuit of FIG. 12 meeting this load condition is Δfs=96.7 kHz, which is a relatively wide range, with fs=117.6 kHz to 208.3 kHz.
The level of the secondary side direct-current output voltage Eo is varied by change in the level of the alternating input voltage VAC, of course. That is, as the level of the alternating input voltage VAC is increased or decreased, the level of the secondary side direct-current output voltage Eo is similarly increased or decreased.
It can thus be said that the level of the secondary side direct-current output voltage Eo is varied more in dealing with the variation of alternating input voltage in a wide range from the AC 100 V system to the AC 200 V system than in dealing with the variation of the alternating input voltage in a single range of only the AC 100 V system or only the AC 200 V system, for example. To perform constant-voltage control operation by dealing with the thus extended variation in the level of the secondary side direct-current output voltage Eo requires a wider necessary control range obtained by extending the above-mentioned range of 117.6 kHz to 208.3 kHz in a direction of higher frequencies.
However, an upper limit of driving frequency handled by an IC (oscillation and drive circuit 2) for driving the switching device in the present situation is about 200 kHz. In addition, even if an IC is developed which can drive the switching device at high frequencies as mentioned above, the high-frequency driving of the switching device significantly decreases power conversion efficiency, and thus makes it practically impossible to put the power supply circuit to practical use.
It is thus understood that it is very difficult to realize a wide range capability by the configuration of the power supply circuit shown in FIG. 12, for example.
Because of such a situation, when a switching power supply circuit including a resonant converter realizes a wide range-capable operation, the switching power supply circuit employs a configuration for switching the configuration of the primary side switching converter to half-bridge/full-bridge configuration according to the commercial alternating-current power supply input of the AC 100 V system/200 V system, for example. Alternatively, the operation of a rectifier circuit rectifying the alternating input voltage VAC is switched to full-wave rectification/voltage doubler rectification according to the commercial alternating-current power supply input of the AC 100 V system/200 V system.
However, the following problems occur when switching is performed between the circuit configurations for the AC 100 V system and the AC 200 V system.
For example, for such switching according to the level of commercial alternating-current power, a threshold value (for example 150 V) is set for input voltage. Circuit switching is performed to the AC 200 V system when the input voltage exceeds the threshold value, and to the AC 100 V system when the input voltage does not exceed the threshold value. When only such simple switching is performed, however, switching may be performed to the AC 100 V system in response to a temporary decrease in the alternating input voltage due to an instantaneous power interruption or the like during input of the AC 200 V system, for example. Specifically, taking a configuration that switches rectifying operation as an example, there is a possibility that the input of the AC 200 V system is determined to be that of the AC 100 V system, and switching is thus performed to a voltage doubler rectifier circuit, so that a switching device or the like exceeds a withstand voltage thereof and consequently breaks down.
Accordingly, in order to prevent the above-described erroneous operation, an actual circuit detects not only the direct-current input voltage of the main switching converter but also the direct-current input voltage of a converter circuit on a standby power supply side.
However, thus detecting the direct-current input voltage of the converter circuit on the standby power supply side means that for example a comparator IC for comparing the input voltage with a reference voltage is incorporated. This increases the number of parts, and thus contributes to increases in circuit manufacturing cost and circuit board size.
In addition, thus detecting the direct-current input voltage of the converter on the standby power supply side for the purpose of preventing erroneous operation means that the wide range-ready power supply circuit can be actually used in only electronic devices having a standby power supply in addition to a main power supply. That is, electronic devices in which the power supply circuit can be mounted are limited to types having a standby power supply, and thus a range of applications of the power supply circuit is correspondingly narrowed.
The configuration that switches between a half-bridge configuration and a full-bridge configuration requires at least four switching devices for the full-bridge configuration. That is, while only two switching devices are required for half-bridge configuration when the switching is unnecessary, two other switching devices need to be added in this case.
The configuration that switches rectifying operation requires two smoothing capacitors Ci to obtain voltage doubler rectifier operation. That is, one smoothing capacitor Ci needs to be added as compared with a configuration that performs only full-wave rectifier operation.
In these respects, the wide range-ready configurations involving the circuit switching as described above increase circuit manufacturing cost and the size of a power supply circuit board. Of parts forming a power supply circuit, a smoothing capacitor Ci or the like in the configuration that switches rectifying operation, in particular, falls under the category of large parts, thus contributing to further increase in the size of the board.
Another problem of the wide control range of the switching frequency as described above is a degradation in quick response characteristics in stabilizing the secondary side direct-current output voltage Eo.
Some recent electronic devices in particular involve a load condition referred to as a so-called switching load, in which load power is changed instantaneously between a maximum load and no load according to for example on/off operation of various driving parts. The power supply circuit needs to perform constant-voltage control on the secondary side direct-current output voltage in response to the load power thus varied quickly over a wide range.
However, with a wide switching frequency control range as described above, it takes a correspondingly long time to change to a switching frequency necessary for the constant-voltage control dealing with the load varied between a maximum value and a minimum value. That is, constant-voltage control response is slow.